Modern integrated operational amplifiers (op-amps) and instrumentation amplifiers (in-amps) bring numerous advantages to design engineers compared to discrete components. While they offer many innovative, functional, and appealing circuit designs, it’s not uncommon for basic problems to be overlooked during the hurried assembly of circuits. This oversight can lead to circuits failing to perform as intended—or worse, not functioning at all. This article aims to explore some of the most frequent application issues and provide practical solutions.
One of the most prevalent problems in operational amplifier and instrumentation amplifier circuits is the absence of a DC bias current loop in AC-coupled systems. In Figure 1, a capacitor is inserted in series with the non-inverting input for AC coupling, a straightforward approach to isolating the DC component of the input voltage (VIN). This technique is particularly beneficial in high-gain applications, where even minor DC voltages at the input can limit the dynamic range and potentially cause output saturation. However, incorporating AC coupling without providing a DC path for the current at the non-inverting input can create significant challenges.
In practice, the input bias current flows into the coupled capacitor, causing it to charge until it surpasses the common-mode voltage threshold of the amplifier's input circuit or limits the output. Depending on the polarity of the input bias current, the capacitor may charge to a positive or negative voltage relative to the power supply. The closed-loop DC gain of the amplifier amplifies this bias voltage.
This charging process can take a considerable amount of time. For instance, in a FET-input amplifier with a bias current of 1 pA coupled to a 0.1 μF capacitor, the charging rate would be 10 μV/s, leading to a 600 μV drift per minute. Such a slow drift might not be noticeable during typical lab tests using an AC-coupled oscilloscope. However, the circuit will likely fail to meet expectations after several hours. Thus, avoiding this issue entirely is crucial.
Figure 2 demonstrates a simple solution to this common problem. By connecting a resistor between the op-amp input and ground, a DC path for the input bias current is established. To minimize the offset voltage caused by the input bias current, especially in bipolar op-amps, the resistances of R1 should ideally match the parallel resistance of R2 and R3.
However, it’s essential to recognize that this resistor introduces additional noise into the circuit. Consequently, there’s a compromise to be made between the circuit’s input impedance, the size of the input coupling capacitor, and the Johnson noise induced by the resistor. Typical resistor values usually range between 100 kΩ and 1 MΩ.
Similar issues can arise in instrumentation amplifier circuits. Figure 3 illustrates an instrumentation amplifier circuit employing two capacitors for AC coupling without offering a return path for the input bias current. This issue is commonly encountered in both dual-supply (Figure 3a) and single-supply (Figure 3b) instrumentation amplifier circuits.
These problems can also manifest in transformer-coupled amplifier circuits, as depicted in Figure 4. This issue arises if no DC-to-ground loop is provided in the transformer secondary circuit.
Figures 5 and 6 showcase a simple resolution for these circuits. High-resistance resistors (RA and RB) are connected between each input and ground. This is a straightforward and practical solution for dual-supply instrumentation amplifiers.
These resistors provide a discharge path for the input bias current. In the dual-supply example shown in Figure 5, the reference terminals of both inputs are grounded. In the single-supply example (Figure 5b), the reference terminals of the two inputs are either grounded (VCM grounded) or linked to a bias voltage, typically half the maximum input voltage.
The same principle applies to transformer-coupled input circuits (refer to Figure 6), unless the transformer's secondary has a center tap that can be grounded or connected to VCM.
In such circuits, a slight offset voltage error can occur due to mismatches in the two input resistors and/or differences in the input bias currents across the two terminals. To minimize this offset error, a resistor can be connected between the two inputs of the instrumentation amplifier (bridging the two resistors), with its resistance being about 1/10th of the first two resistors (though still large compared to the differential source impedance).
Figure 6 highlights the correct instrumentation amplifier transformer input coupling method.
Properly Providing Reference Voltages for Instrumentation Amplifiers, Op-Amps, and ADCs
Figure 7 presents a single-supply circuit where an instrumentation amplifier drives an analog-to-digital converter (ADC) with a single-ended input. The amplifier's reference voltage provides a bias voltage for a zero differential input, while the ADC's reference voltage serves as a scaling factor. A simple RC low-pass anti-aliasing filter is typically placed between the output of the instrumentation amplifier and the input of the ADC to reduce out-of-band noise.
Design engineers often seek to use straightforward methods like resistor dividers to provide reference voltages for instrumentation amplifiers and ADCs. However, this can lead to significant errors when using certain types of instrumentation amplifiers.
Figure 8 illustrates the incorrect use of a simple resistor divider to directly drive the reference voltage of a three-op-amp instrumentation amplifier.
For example, the popular instrumentation amplifier design configuration employs the three-op-amp structure shown above. The total signal gain is:
\[ \text{Gain} = \left(1 + \frac{R2}{R1}\right) \cdot \left(1 + \frac{R4}{R3}\right) \]
The gain at the reference voltage input is 1 (if input from a low-impedance voltage source). However, in the circuit shown above, the reference input pin of the instrumentation amplifier is directly connected to a simple voltage divider. This alters the symmetry of the subtractor circuit and the voltage divider ratio of the voltage divider. It also reduces the common-mode rejection ratio of the instrumentation amplifier and its gain accuracy. However, if R4 is connected, the equivalent resistance of the resistor becomes smaller, and the reduced resistance value is equal to the resistance value (50 kΩ) seen from the two parallel branches of the voltage divider. The circuit behaves as a low-impedance voltage source that is half the supply voltage applied to the original value of R4, and the accuracy of the subtractor circuit remains unchanged.
This method cannot be used if the instrumentation amplifier is in a closed single package (one IC). Additionally, consider that the temperature coefficient of the divider resistor should match the resistor in R4 and the subtractor. Finally, the reference voltage will not be adjustable. On the other hand, if you attempt to reduce the resistance of the voltage divider resistor, the increased resistance is negligible, which increases the power supply current consumption and the power consumption of the circuit. In any case, this clumsy method is not a good design.
Figure 9 shows a better solution by adding a low-power op-amp buffer between the voltage divider and the instrumentation amplifier's reference voltage input. This eliminates the impedance matching and temperature coefficient matching issues, and it makes adjusting the reference voltage straightforward.
Proper PSR Performance When Using Voltage Dividers to Supply Reference Voltages from the Supply Voltage
A frequently overlooked problem is that any noise, transients, or drift in the supply voltage VS will be directly applied to the output via the reference input as a function of the voltage division ratio. Practical solutions include bypass filtering and even reference voltages generated using precision reference voltage ICs, such as the ADR121, rather than VS divider.
This consideration is critical when designing circuits with instrumentation amplifiers and operational amplifiers. Supply voltage suppression techniques are used to isolate the amplifier from hum, noise, and any transient voltage variations in its supply voltage. This is vital because many actual circuits are included, connected, or exist in environments that provide only non-ideal supply voltages. Furthermore, the AC signal in the power line can feedback into the circuit to be amplified and, under appropriate conditions, cause parasitic oscillations.
Modern op-amps and instrumentation amplifiers offer relatively low-frequency supply voltage rejection (PSR) capabilities as part of their design. Most engineers take this for granted. Many modern op-amps and instrumentation amplifiers have PSR specifications above 80 to 100 dB, which can attenuate the effects of supply voltage variations to 1/10,000 to 1/100,000. Even a 40 dB PSR amplifier provides a 1/100 rejection of the power supply. However, high-frequency bypass capacitors (as shown in Figures 1-7) are always required and often play a key role.
Additionally, when a design engineer uses a simple supply voltage resistor divider and employs an op-amp buffer to provide a reference voltage for the instrumentation amplifier, any change in the supply voltage passes directly through the circuit to the output stage of the instrumentation amplifier without attenuation. Thus, unless a low-pass filter is provided, the IC typically experiences a significant loss in PSR performance.
In Figure 10, a large capacitor is added to the output of the voltage divider to filter out variations in the supply voltage and ensure PSR performance. The -3 dB pole of the filter is determined by the parallel connection of resistor R1/R2 and capacitor C1. The -3 dB pole should be set at 1/10 of the lowest useful frequency.
A trial value of CF is capable of providing a –3 dB pole frequency of approximately 0.03 Hz. A small capacitor (0.01 μF) connected across R3 minimizes resistor noise.
This filter takes time to charge. According to the trial value, the rise time of the reference input should be several times the time constant (here T=R3Cf= 5 s), or 10~15 seconds.
The circuit in Figure 11 has been further improved. Here, the op-amp buffer acts as an active filter that allows the use of capacitors with much smaller capacitance values to decouple the same large supply. Additionally, an active filter can be used to increase the Q value to speed up the on-time.
Test Results: Using the component values shown above, applying a 12 V supply voltage to filter the 6 V reference of the instrumentation amplifier. Setting the gain of the instrumentation amplifier to 1 and modulating the 12 V supply with a frequency-changing 1 VP-P sinusoidal signal. Under such conditions, as the frequency decreases, it is reduced to approximately 8 Hz, and we do not see the AC signal on the oscilloscope. When applying a low-amplitude input signal to an instrumentation amplifier, the circuit's test supply voltage range is 4 V to 25 V or higher. The on-time of the circuit is approximately 2 seconds.
Decoupling of Single-Supply Operational Amplifier Circuits
Finally, a single-supply op-amp circuit needs to bias the common-mode input voltage amplitude to control the positive and negative swings of the AC signal. When the bias voltage is supplied from the supply voltage using a voltage divider, proper decoupling is required to ensure PSR performance.
A common but incorrect method is to provide a VS/2 to the non-inverting input of the op-amp using a 100 kΩ/100 kΩ resistor divider (plus a 0.1 μF bypass capacitor). Decoupling the supply with such a small capacitance value is usually insufficient because the pole is only 32 Hz. The circuit is unstable ("low-frequency oscillation"), especially when driving inductive loads.
Figures 12 (inverting input) and 13 (non-inverting input) show the VS/2 bias circuit that achieves the best decoupling results. In both cases, the bias voltage is applied to the non-inverting input, fed back to the inverting input to ensure the same bias voltage, and the unit DC gain is also biased by the same output voltage. Coupling capacitor C1 reduces the low-frequency gain from BW3 to unity gain.
As shown in the figure above, a good rule of thumb when using a 100 kΩ/100 kΩ resistor divider is that the minimum C2 should be 10 μF to achieve a –3 dB cutoff frequency of 0.3 Hz. And 100 μF (0.03 Hz) is actually sufficient for all circuits.
Figure 13 shows the correct decoupling circuit for a single-supply inverting input amplifier with a gain of –R2/R1.
This comprehensive explanation covers various practical solutions to common issues encountered in operational amplifier and instrumentation amplifier circuits, ensuring reliable performance under diverse conditions.
Wireless CPE
What is 5G CPE?
Definition of 5G CPE
CPE stands for Customer Premise Equipment. The so-called front end refers to the equipment in front of the customer's terminal equipment. When we use Wi-Fi, if the distance is far, or there are more rooms, it is easy to appear signal blind spots, resulting in mobile phones or ipads or computers can not receive Wi-Fi signals. The CPE can relay the Wi-Fi signal twice to extend the coverage of Wi-Fi.
What are the benefits of CPE?
Through the following comparison table, it is not difficult to understand the technical advantages of CPE products:
* Currently, the global 5G FWA service is mainly in the Sub-6GHz band, with only the United States and Italy supporting the millimeter wave band.
* 5G CPE integrates the low cost of Wi-Fi and the large bandwidth of 5G, combining the advantages of the two to form a strong complement to traditional fiber broadband.
The relationship between 5G, FWA and CPE
It can be said that FWA (Fixed Wireless Access) will be the most down-to-earth application of 5G technology. FWA business plays a key role in enabling "connecting the unconnected." FWA is a low-cost, easy-to-deploy flexible broadband solution. Compared with wired access technology, FWA has been an ideal choice for deploying broadband in many countries and regions because it does not need to obtain rights of way, dig trenches and bury cables, and drill holes through walls. The development of 5G technology is further promoting the development of FWA.
FWA services (including 4G and 5G) have reached 100 million users. FWA is no longer a niche service; The FWA industry as a whole has been supported by numerous suppliers. Why is that? In the 5G era, 5G CPE receives 5G signals from operator base stations and then converts them into Wi-Fi signals or wired signals to allow more local devices to get online. For operators, the initial user penetration rate of 5G is low, and the investment is difficult to realize quickly; The CPE business can use the idle network to increase revenue for operators, so major operators vigorously promote the development of 5G CPE.
FWA services can be used for both home (To C) and business (To B), and customers have different requirements for CPE devices when using FWA services in different application environments, resulting in consumer grade 5G CPE and industrial grade 5G CPE (similar to home routers and industrial routers).
In 2020, the global market size of 5G CPE will reach 3 million units, and it is expected that in the next five years, the market size of 5G CPE will maintain a compound growth rate of more than 100%, reaching 120 million units in 2025, with a market value of 60 billion yuan. As an important market for 5G CPE, China's 5G CPE market size will reach 1.5 million units in 2020 and is expected to reach 80 million units in 2025, with a market value of 27 billion yuan.
The difference between 5G CPE and other devices
CPE can support a large number of mobile terminals that access the Internet at the same time, and the device can be directly inserted with a SIM card to receive mobile signals. CPE can be widely used in rural areas, cities, hospitals, units, factories, communities and other wireless network access, can save the cost of laying wired networks.
A Router is a hardware device that connects two or more networks, acts as a gateway between networks, and is the main node device of the Internet. Routers use routes to determine the forwarding of data. If it is a home router, it does not support a SIM card slot, and can only receive signals by connecting to optical fiber or cable and then convert it into WI-FI to provide a certain number (several) of terminal devices to surf the Internet.
Industrial 5G CPE is equivalent to 5G industrial routers, and the technology of the two is not very different. On the one hand, the industrial 5G CPE converts 5G network signals into WiFi signals for transmission, and on the other hand, the data received by the WiFi network is converted into 5G network signals for uploading. In addition, industrial 5G CPE generally supports routing functions.
5G CPE trends
According to a research report, after evaluating the products of some mainstream 5G CPE suppliers, many institutions believe that the development of 5G CPE products will continue in two aspects: one is to support mmWave and Sub-6 GHz at the same time; Second, the design will pay attention to humanized operation and installation. The industry development trend will accelerate the demand for 5G in the medical, education and manufacturing industries due to the epidemic, and 5G FWA will promote global 5G CPE shipments.
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